Wireless devices are equipped with antennas which radiate a signal having a physical orientation called polarization. Polarization can be horizontal, vertical, or circular, depending on the antenna's orientation. When travelling across unobstructed environments, the signal maintains its polarization and reaches the receiving antenna unchanged. If the receiving antenna exhibits the same polarization as the incoming signal, then a high signal power is obtained at the receiver and a good data rate ensues. In indoor and urban environments however, the random presence of numerous reflective and diffractive objects leads to a change in the signal's polarization (from horizontal to vertical or from vertical to horizontal). If the receiving antenna is not oriented in the same manner as the incoming signal's unknown polarization, then a lower signal power is obtained. An antenna which is capable of simultaneously receiving both horizontal and vertical polarizations or is able to switch between polarizations is ideally suited for highly reflective indoor environments.
To combat reflective and diffractive effects, one approach is to use circularly-polarized (CP) antennas. First, CP antennas reduce polarization losses because the signal is transmitted and/or received in all planes. Secondly, due to its radial rotation, a CP signal better propagates through obstructions, such as walls (e.g. reinforced concrete with vertical metallic stubs), than linearly-polarized signals, which increases the range and coverage area. Thirdly, CP antennas alleviate the effects of multi-paths, reducing small-scale fading.
Early work on serially-fed CP arrays involved waveguide and planar implementations. In a publication by K. Sakakibara et al., published in IEEE Trans. Vehic. Techn., vol. 48, no. 1, pp. 1-7, in January 1999, a pair of orthogonal slots are etched on the top surface of a waveguide, with each pair being separated by a quarter-guided wavelength λ_g/4 to achieve CP. However, this array cannot scan the full-space, and only one CP (right or left) is obtained when the signal is injected from either end.
The composite right/left-handed (CRLH) leaky-wave antenna (LWA) is considered as a serially-fed array. It is advantageous over corporate-fed arrays due to the absence of a complex feeding network, making it more compact, less lossy and less costly, as reported in a publication by L. Lui et al., published in Electron. Lett., vol. 38, no. 23, pp. 1414-1416, in November 2002.
In a publication by M. Hashemi and T. Itoh, published in IEEE Intern. Conf. Wirel. Infor. Techn. and Systems, Honolulu, Hi., on Aug. 28, 2010, a hybrid coupler is connected to two orthogonal CRLH LWAs to achieve CP. The antennas can scan the full-space with both polarizations obtained from a single end. However, this configuration greatly increases the structure's form factor and is not conducive to array implementations. Also, in a publication by Y. Dong and T. Itoh, published in Proc. of APMC, Yokohama, Japan, on Dec. 7-10, 2010, a hybrid coupler is connected to two co-planar substrate-integrated CRLH LWAs each having oppositely slanted slots for CP. This antenna can also scan the full-space, and provide both polarizations from a single end. However, it is not adaptable for single-frequency electronic-scanning.
The CRLH LWA has been receiving increased attention due to its many advantages, as explained by L. Liu, C. Caloz, and T. Itoh in a publication entitled “Dominant mode (DM) leaky-wave antenna with backfire-to-endfire scanning capability”, published in Electron. Letters, vol. 38, no. 23, pp. 1414-1416, on November 2002. Fundamentally, the CRLH LWA is a beam-scanning antenna similar to phased arrays, however without the bulky and lossy feeding network and without the lossy and costly phase shifters. This ideally allows CRLH LWAs to be used in various beam-steering communication applications such as radars, satellite communications, and Wireless Fidelity (WiFi).
CRLH LWAs have undergone many improvements since their discovery, as reported in a publication from M. Hashemi and T. Itoh, entitled “Evolution of composite right/left-handed leaky-wave antennas,” published in Proc. of the IEEE, vol. 99, no. 10, pp. 1746-1754, in October 2011. One such improvement includes a recently proposed CRLH which exhibits dual-polarization, described in a publication from M. R. M. Hashemi and T. Itoh, entitled “Dual-mode leaky-wave excitation in symmetric composite right/left-handed structure with center vias”, published in Proceedings. Institute of Electrical and Electronics Engineer (IEEE) MTT-S Int. Microwave Symposium, Anaheim, Calif., in May 2010, pp. 9-12, hereinafter “Hashemi 2010”, from which FIG. 1 shows a dual-polarized CRLH LWA. The CRLH LWA 1 of Hashemi 2010 is composed of two symmetric CRLH LWAs 2 and 3 sharing a common via 6, with each CRLH LWA being composed of series inter-digital capacitors 5 and shunt stub inductors 4. By exciting the structure at V1 and V2 with a common-mode (CM) signal and a differential-mode (DM) signal, the CRLH LWA radiates in horizontal-polarization (along x) and in vertical-polarization signal (along y), respectively. However, this dual-polarization CRLH LWA suffers from a major drawback which is the unbalanced behavior of the common and differential modes. This unbalanced behavior leads to undesired effects. Under separate common-mode and differential-mode excitations, the dual-polarized CRLH LWA's unit cell (UC) can be modeled using two different electrical circuits.
FIG. 2 shows the circuit diagram for the dual-polarized CRLH LWA of Hashemi 2010 using a shared via. More specifically, FIG. 2 provides a close-up of the dual-polarized CRLH LWA's unit cell 7. Under separate CM and DM excitations, the UC 7 can be modeled using two different electrical circuits. As shown in a lower part of FIG. 2, the common-mode and differential-mode circuit diagrams are similar except that two inductors, which model the shared via 6 with a value of Lvia/2, are missing from the differential mode circuit diagram. This is due to the differential excitation inducing a virtual short circuit in the middle of the structure which shorts out the via 6.
FIG. 3 shows the modal reflection (S11) and transmission (S21) responses for common-mode (Scc) and differential-mode (Sdd) excitations for the dual-polarized CRLH LWA of Hashemi 2010 using a shared via. In the Figure, S11 refers to a reflection coefficient (from a port 1 back to the same port 1) while S21 refers to a transmission coefficient (from a port 1 to another port 2). FIG. 4 shows the modal transmission (S21) phase response for common-mode (Scc) and differential-mode (Sdd) excitations for the dual-polarized CRLH LWA of Hashemi 2010 using a shared via. The effect of having two different circuit behaviors under common-mode and differential-mode excitations is shown in FIGS. 3 and 4, where the modal (common and differential) scattering parameters' amplitude and phase responses are shown, respectively. As can be seen, the common-mode and differential-mode S-parameters (Scc and Sdd) are dissimilar in both magnitude and phase. Ultimately, as a result of the different modal responses, the dual-polarized CRLH LWA radiates two different beams (in shape and pointing angle) under common-mode excitation (for horizontal polarization) and differential-mode excitation (for vertical polarization), thus making the proposed dual-polarized CRLH LWA of Hashemi 2010 impractical.
There is therefore a need for a polarization-diverse CRLH LWA and for an antenna system employing the polarization-diverse CRLH LWA.